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Abstract: A novel Inset fed microstrip patch antenna is designed on defected ground structure (DGS) to reduce the higher order harmonics and cross polarized radiations. Square rings, and square shaped slots are placed on the ground plane of the microstrip patch antenna to get the desired operation. These annular ring and arc DGS appears to be highly efficient in terms of suppressing the cross polarization. Relative suppression of radiated field is observed by placing and without placing the defected ground structures. The stop band property exhibited by the DGS is used to filter out the harmonics. The current model successfully reduced the DGS size and by comparing with the well known design, size reduction of 20% is achieved. Instead of normal square patch, a slotted aperture patch model is considered in the current design to reduce the overall size of the antenna.
Abstract—It has been shown that it is possible to get a good estimation of the location of the largest centers of reflection causing ripple in the quiet zone using spherical near-field scanning of the quiet zone in combination with back projection to far-field. This method however, suffers from poor resolution at lower frequencies making it hard to distinguish small contributions from the main beam if they are closely spaced. For this purpose the CLEAN algorithm has been adapted and is presented here.
Abstract—The measurement geometry and data processing techniques employed in spherical near-field (SNF) antenna measurements naturally quantify the directivity of an antenna under test. Computing antenna gain from these measurements requires additional information and processing. Equations that can be used to calculate the magnitude of antenna gain from spherical near-field measurements are provided in seminal SNF references, but equations that describe how to calculate the complex gain voltage of an antenna with spherical near-field methods have been largely absent in the literature. This paper presents equations that may be used to calculate the complex gain voltage of an antenna using the gain substitution method in a spherical near-field test range. The equations are presented in a more generalized format than previously seen, and will show how to use a combination of data collected in the near-field with data transformed to the far-field to calculate the gain voltage. Practical examples are provided for determining gain voltage using a single measurement set-up or multiple measurement set-ups, including a method to calculate gain voltage of each port of a multi-port antenna requiring only a single full sphere measurement of the standard gain antenna.
I. INTRUDUCTION In the heating process of microwave absorbers under incident electromagnetic waves, two disciplines of physics are intertwined, i.e., electromagnetic waves behavior governed by Maxwell’s equations and heat transfer process dictated by laws of thermodynamics. The power density in the absorbers due to the electromagnetic .eld is given by p= s|E|2 =2po0 o ' f|E|2 (1) where, E is the total electric .eld (V/m) in the material, s is electrical conductivity of the material (S/m), o0 is the free space permittivity (8.854 × 10-12 F/m), o' is the imaginary part of the relative dielectric constant, and f is the frequency in Hz. This is point form of the Joule’s law, and is well understood by RF engineers. The EM behavior of the polyurethane absorbers can be numerically computed. The EM .eld acts as the heating source, and its distribution in the absorber can provide a good indication on the locations of hot spots. Polyurethane foam is an excellent insulator, so the conductive heat loss may be minimal. The heat exchanges can be reasonably described by radiation and convection transfers. Radiation takes place in the form of EM wave, mainly in the infrared region. The net power transferred from a body to the surroundings is described by Stefan-Boltzmann’s law [1], prad = osA(T4 -T04 ) (2) where A is the surface area, T is the surface temperature of the radiation body in K, and T0 is the ambient temperature in K. Unfortunately, the conventional symbols used in heat transfer s and o are not the same as those in Eq. (1). s here is the emissivity or emission coef.cient, and is de.ned as the ratio of the actual radiation emitted and the radiation that would be from a black body. o in Eq. (2) is the Stefan-Boltzmann constant (5.67 × 10-8 W/m2 K4 ). The context in the paper should make it clear which symbols the authors are referring to. Otherwise, we will make explicit references. The convective heat transfer is due to the motion of air surrounding the absorbers. Two forms can take place, naturally or by forced air. The relationship is described by Newton’s law of cooling [1]: pconv = hA(T -T0 ) (3) where h is the convection heat transfer coef.cient in (W/m-2 K-1 ). h is often treated as a constant, although it can be a function of the temperature. Eq. (3) assumes that the ambient air is abundant, and is taken to be constant. This is a reasonable assumption, because the heating is typically con.ned to a small localized area in a relatively large anechoic chamber. Combining the two mechanisms of heat transfer, the total heat loss is given by p= osA(T4 -T4 )+ hA(T -T0 ) (4) 0 It is possible to solve for the temperatures from coupled Maxwell’s and heat transfer equations. Realistic results require accurate electrical and thermal properties of the materials. It is often a non-trivial process to obtain the material properties in and of itself. Careful validation is warranted before we can have full con.dence in the results. In this paper, we adopt a measurement approach instead. We conduct a series of experiments to measure the temperature both on the surface of the absorbers using an infrared imaging camera, and internally using thermocouple probes inserted into the absorbers. Temperature pro.les versus applied E .eld are experimentally established. From the measured data, we curve .t to Eq. (4) or other mathematical functions. These functions are useful to calculate results at other .eld levels, e.g., extrapolating to a higher .eld where measurement results cannot be readily obtained. II. FIELD DISTRIBUTION INSIDE THE ABSORBERS Numerical analysis was performed using Ansys HFSS, a commercially available Finite Elements software package. As it was described in [2], symmetry is taken advantage of, so only one quarter of the pyramidal absorber is solved. The quarter pyramid is located inside a square cross section prism that bounds the computational domain. The structure is fed using a port located on the top of the geometry and the side boundaries of the domain are set as perfect electric conductor (PEC) or perfect magnetic conductor (PMC). The base is modeled as PEC. This is exactly the same approach taken in [2]. The structure of a CRV-23PCL-4 is analyzed at 12.4 GHz, the same frequency as used in the measurements. The resulting .eld is extracted at one plane. The plane is one of the two orthogonal planes that cut the pyramid in 4 sections. Fig. 1 shows the .eld distribution at 12.4 GHz. The curvature of the absorber pro.le has been added for clarity. The results are an approximation. The permittivity of the material is assumed to be fairly constant from 6 GHz to 12 GHz. The purpose of the numerical analysis is to check the expected .eld distribution in the pyramid, which we can use to compare with the infrared (IR) images of the absorbers taken during the measurements. Fig. 1. Electric Field distribution at 12.4 GHz The .eld distribution data shows that most of the .eld exists on the upper third of the pyramid. It also shows that there is a region of high .eld existing in the valleys between the pyramids. The surface temperature pro.le from the IR pictures shows that this is an real phenomena. On the other hand, the .eld is higher at the very tip of the absorber. Measurements from the IR images seem to contradict this result. This can be explained. Since the tip is smaller, it cools faster to the surrounding ambient temperature. III. EXPERIMENTAL SETUP AND DATA Experiments were performed on ETS-Lindgren CRV-23PCL-8, and CRV-23PCL-4 absorbers at 12.4 GHz. Both types are 23” long from tips to bases. A piece has a base size of 2’ × 2’. A CRV-23PCL-8 piece consists of 8×8=64 pyramids, whereas a CRV-23PCL-4 piece consists of 4×4=16 pyramids. The two types are designed to have similar RF performances, but the CRV-23PCL-8 is made of slender pyramids to facilitate better heat transfers to the surroundings [2]. The absorbers are mounted on a particle board with metallic backings, and are placed in front a Ku band horn antenna with a circular aperture (the gain is approximately 20 dBi). A 300W ampli.er is used, and the power to the antenna is monitored through a 40 dB directional coupler connected to a power meter. The test setup is shown in Fig. 2. The ambient temperature is at 23.C. Fig. 2. Test setup using a conical horn antenna to illuminate the absorbers As a .rst step, a 200 V/m .eld is generated by leveling to a calibrated electric .eld probe. The distance from the probe to the antenna is 30”. At this distance, near .eld coupling is assumed neglegible, and the incident wave uniform (numerical simulation also validated these assumptions). The power needed to generate 200 V/m .eld is recorded. Next, the .eld probe is replaced with the absorbers under test. The tips of the absorbers are placed at the same distance (30”) from the antenna. Other .eld strengths can be leveled by scaling from the power for 200 V/m. A. Surface Temperature Figs. 3 and 4 show two examples of the infrared images taken after the temperature reached equilibrium under a constant 700 V/m CW at f=12.4 GHz for the two types of absorbers described earlier. There is no forced air.ow during the measurement. Table 1 summarizes the resulting temperatures on the absorber surfaces at different .eld levels. Tests were performed on two .nishes of otherwise identical CRV-23PCL-8 absorbers, i.e., fully covered with rubberized paint, or with latex paint. The data indicates that the paint has minimal effects on absorber temperatures. Table 1 also lists data for the wider CRV-23PCL-4 absorbers (with latex paint). B. Internal Temperature of the Absorber recorded by Thermocouples Three thermocouples are inserted in the CRV-23PCL-8 which are painted with rubberized coating. They are inserted at distances of 4”, 6”, and 8” from the tip of the pyramid, as illustrated in Fig. 5. Fig. 6 shows the temperatures measured by the three sensors. The temperatures at 8” from the tip are consistently higher than at other locations. There is a gap in the data at 700 V/m because RF power was turned off brie.y. Internal temperature reached 115 .C under 1.7 kW/m2 Fig. 3. Infrared camera image for incident electric .eld of 700 V/m. The absorber is the slender CRV-23PCL-8. Fig. 4. Infrared camera image for incident electric .eld of 700 V/m. The absorber is the wider CRV-23PCL-4. (800 V/m). Since the maximum allowed temperature for the polyurethane foam material is 125 .C, the incident power density is recommended to stay less than 1.7 kW/m2 for CRV-23PCL-8 absorbers mounted vertically and with natural convection in a 23.C room. After the temperature reached equilibrium under 800 V/m, additional air.ow was introduced by turning on a 6” diameter fan at 45” in front of the absorbers. The air.ow rate was measured to be approximately 80 ft/min at this distance. Note that this is a rather moderate air.ow, which can arise naturally from air-conditioning vents in a chamber. As shown in Fig. 6, the internal temperature quickly dropped to 102.C from 115.C. TABLE I MAXIMUM SURFACE TEMPERATURE RECORDED BY THE IR CAMERA (AT EQUILIBRIUM). T0 =23. C. E Power CRV-23PCL-8 CRV-23PCL-8 CRV-23PCL-4 (V/m) Density rubberized latex (. C) rubberized (kW/m2 ) (. C) (. C) 200 0.11 24 300 0.24 28 360 0.34 30 400 0.42 35 36 43 500 0.66 41 50 600 0.95 54 67 700 1.30 63 82
Abstract—The optimal allocation of coaxial-slot antennas for hyperthermia treatment has been determined by using the p-center location problem, which is an operations research problem. Some examples of various p-center solutions in a square are demonstrated. As a result, we clarified that the optimal coordinates and minimum number of antennas can be determined once the area of the tumor and the medical treatment radius of the coaxial-slot antenna are determined. Moreover, the specific absorption rate distribution inside the biological tissue-equivalent phantom has been calculated using the finite-difference time-domain method.
Abstract—A new technique is presented to measure the permittivity and permeability of conductor-backed magnetic absorbing materials using a triaxial probe. The probe consists of two coaxial transmission lines that share an aperture in a conducting flange, which is placed against the sample creating a two-port network. By measuring the reflection coefficients at each port and the transmission between the ports, the material parameters may be determined. This paper describes the technique and provides a theoretical method for computing the S-parameters of the triaxial system. Experimental implementation of the system is still under study.
The need to maintain very low observability, along with the need to manipulate the model through a large range of motion, result in a challenging set of problems. These have been effectively addressed over decades of RCS equipment design. In recent years however, RCS applications have become much more demanding. Models are ever larger and heavier, with length exceeding 150 feet, and with weight up to 50,000 lbs. Required accuracy with some applications has increased to ±0.01°, an increase of 67% as compared to legacy values. MI Technologies has developed products that significantly expand the structural and operational envelopes of rotator/pylon systems to meet the demand for higher performance. This paper presents the various challenges encountered in RCS Rotator and Pylon design, and the innovative solutions that have arisen from recent engineering efforts.
In this paper we describe the NGS calibration facility and calibration hardware, and discuss the motivation for providing calibration services. We provide the time-difference, single-difference carrier phase observable models and estimation strategy currently used to generate NGS absolute calibrations. Calibration examples are provided in the accompanying poster presentation.
Modern antenna measurement systems require high dynamic range to accommodate a wide variety of high gain and low gain antenna testing. Network analyzers utilized as the baseline instrument in these systems (e.g., Agilent PNA series), typically offer more than adequate dynamic range. Of course, this is reduced to some extent by cable losses and other components in the transmission and reception paths. However, the capability for achieving the required dynamic range still is adequate in most RF configurations given the proper choice of mixers, amplifiers, etc. in the layout. However, often the achievable dynamic range is limited relative to the calculated values; this is quite often a result of leakage signals within the RF paths. The primary culprits of the leakage are typically rotary joint leakage in positioners and direct leakage into cable connectors. Various tests are required to isolate the leakage signals and to quantify them. This paper will identify the techniques used to isolate these signals and offer solutions to minimize the leakage and thus optimize dynamic range. Recent testing has shown that leakage signals may be as much as 10-20 dB or more above the achievable noise floor when commercially available rotary joints and multiple cables are utilized. These issues will be addressed in the paper.
Phase variation = +/-10 deg. 18 to 40 GHz Phase variation = +/-20 deg. 40 to 110 GHz Cross Polarization = -30 dB III. MAXIMUM AVAILABLE SPACE Consistency of performance across a waveguide band levies demands on compact range feeds. Because of the constraint of the room size, the design starts with determining the maximum space available for the This paper addresses a recent compact range development by MI reflector. The next step will be to determine the combination of Technologies that achieves desired extended low frequency and reflector body and edge treatment size within that space to millimeter wave performance (1 to 110GHz) while maintaining a deliver the desired performance. To determine the space cost effective reflector size and a small range footprint. The paper available for the reflector a chamber layout analysis is will explore the conventional rule-of-thumb relationships performed. Appropriate absorber is selected and, allowing for between feed, reflector, edge treatments and range geometries an air gap of at least 2 wavelengths at the lowest desired while contrasting them to the resultant design. The paper will frequency between the absorber and the reflector, and allowing highlight an impressive new family of compact range feeds and advancements in cost effectively achieving a superior reflector height for the compact range feed positioner yields the surface. allowable reflector dimensions to be 194 inches high and 222 inches wide as shown in Table 2. The combination of reflector
Abstract— Plasma antennas have more degrees of freedom then metal antennas making their applications have enormous possibilities. Plasma antennas use partially or fully ionized gas as the conducting medium instead of metal to create an antenna. The advantages of plasma antennas are that they are highly reconfigurable and can be turned on and off. Haleakala Research and Development, Inc. has done theoretical work, experiments, and built prototypes on plasma antennas, plasma waveguides, and plasma frequency selective surfaces. This research and development focused on reducing the power (to date this is less than 5 watts per tube and going down) required to ionize a plasma tube with higher plasma densities and frequencies, plasma antenna nesting, co-site interference reduction, thermal noise reduction, and the development of the smart plasma antenna. Haleakala Research and Development, Inc. has built and tested plasma antennas from 30 MHz to 50 GHz. At the higher frequencies, plasma antennas have lower thermal noise than metal antennas and the thermal noise of plasma antennas decreases with the operating frequency of the plasma antenna making them ideal for satellite antennas. Plasma satellite antennas can be made conformal with a surface and give the performance of a parabolic dish antenna. This is true because beam steering and focusing can be done by varying the plasma density from one tube of plasma compared to the next. With this design, plasma satellite antennas can be operated in the reflective or refractive mode. High powered plasma antennas have been built as a possible directed energy applications. Alexeff and Anderson and [1]-[2] Anderson and Alexeff [3] have done theory, experiments, and have built prototype plasma antennas. Anderson [4] wrote a comprehensive book on plasma antennas.
Abstract—In this paper, a novel feeding method of microstrip line array is presented, with which every line array can be fed from the side part instead of from the center part .At the same time a novel compact slot patch antenna loaded with a pair of spirals is proposed, which is 66% smaller than the conventional half-wave patch antenna. The simulated and measured radiation patterns at center frequency are both presented. The -18dB side-lobe level for the sum pattern and -37dB null depth for the difference pattern have achieved in the experiment.
Abstract—Nowadays, radar retro-reflector has been widely applied as a decoy, to seduce an incoming assault away from the target, or towards a less vulnerable part of it to communication systems and remote identification as their characteristics of low-profile, low-cost and Radar Cross Section(RCS) enhancement. A passive retro-reflector is a device which can be used to be reflected most of the energy incident upon it in the direction of the in-going wave. The Luneberg lens and a sphere are widely used as their self characteristics. In this paper one of the retro-reflector, is paid more attention as time goes by, is introduced. The retroreflector is consist of patch antenna arrays and feeding system and can be defined as Retro-directive arrays (RDA). It has a very simple structure and can focus outgoing waves back at the direction of incident waves. The character of the re-radiation pattern affected by the size and type of patch and width and length of feeding network related are optimized by the HFSS. The final results are validated experimentally.
Abstract—When antennas are measured in echoic environments, there is usually a need to process the measured data in order to remove multipath contributions. Traditional measurement set-ups involving a single probe antenna provide limited information for the purpose of this separation. A multi-probe measurement technique whereby the AUT is measured with two sets of probe antennas is presented. One set of probe antennas are oriented such that they radiate mainly toward the AUT and the other set radiate away from it. This measurement technique allows for the separation of the direct AUT contribution from the multipath contributions. The acquired data is processed using a well-suited near-.eld far-.eld transformation algorithm with the echo sources considered as they were independent sources. The performance of the measurement technique is also evaluated for traditional spherical mode near-.eld far-.eld transformation whereby both incoming and outgoing spherical waves are considered. The results show a substantial improvement in the obtained far-.eld patterns when compared with non-compensated far-.eld results.
Abstract—This contribution introduces a technique that offers the possibility to reconstruct all scattering parameters of a given sample based on re.ection measurements only. Therefore, it is possible to obtain the transmission properties of a given device under test in a measurement setup which does not allow to perform transmission measurements. Thus, the new method combines the advantage of a two port measurement, which yields all four scattering parameters, with the more compact setup of a one port measurement. Free space material measurements are performed to validate the new approach.
Abstract— Compact ranges are ideal settings for collecting low-RCS measurement data at high pulse rates. However, until recently, two operating constraints have limited the efficiency of instrumentation radar systems in this setting: (1) system delays limiting Pulse Repetition Frequency (PRF) and (2) fixed integration across frequency resulting in more time spent on certain frequencies than required. In this paper, we demonstrate the capability to significantly increase data throughput by using a Burst-Mode to increase the usable PRF and a frequency table editing mode to vary integration levels across the frequency bandwidth. A major factor in the choice of PRF for a specific application is system hardware delays. We describe the use of a Burst-Mode of operation in the MkVe Radar to reduce delays caused by physical layout of the instrumentation hardware. Burst-Mode essentially removes setup time in the system, reducing the time between pulses to the roundtrip time of flight from the antenna to the target. Most pulsed-IF instrumentation radar users fix the coherent integration level for the entire measurement waveform, even though the set level of integration may not be required at all frequencies to achieve the desired sensitivity. We describe the use of a frequency table Parameter Editor Mode in the MkVe that allows the integration level to vary for each step in the waveform. We demonstrate the use of both methods to reduce data collection time by a factor of seven using a MkVe Radar installed in a compact range.