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In array signal processing; i.e. direction of arrival estimation, digital beam forming/ nulling one needs to know the antenna array manifold (relative gain and phase of individual antenna elements) as well as the relative response of RF front end of the various channels. Here the RF front end is defined as LNA, filters, various down convertors and A/Ds. Since RF front end response is highly dependent on the physical environment, in general, nearly real time calibration of the RF front end is carried out. To accomplish this, a pilot signal is injected in various channels or a strong signal is received from a known direction. The received pilot is isolated, and the isolated signal is processed for relative calibration of the various front ends. One simple processing technique is to divide the signal received at a given frequency by the channel of interest with the signal received by the reference channel at that frequency. This technique works fine when the pilot signal has high SNR (20 dB or so). For low SNR, the front end calibration will have large variance and also bias, which is undesired. One can also cross correlate the signal received by the two channels at the frequency of interest, and normalize the cross correlation with the autocorrelation of the signal received by the reference channel. We will show that for low SNR of the pilot signal, this approach can lead to bias in the estimated relative magnitude. We will also present a novel approach that leads to unbiased estimate with small variance. The approach is based on signal space idea. At the given frequency, we generate a covariance matrix that contains the correlation between the signals received by various channels. Next, the principle eigenvector (corresponding to the largest eigenvalue) of the covariance matrix is calculated. The eigenvector is adjusted such that its element corresponding to the reference channel is unity. Then the other elements of the eigenvector yield the relative response of the front end of various channels. We have applied the suggested approach to real world data with very good results. Some examples are discussed in the paper.
Abstract – This paper introduces a method for calibrating the gain and the frequency dependent phase center locations of Log Periodic Dipole Arrays (LPDAs). The method builds upon the three antenna method, but is conducted over a PEC ground plane in an Open Area Test Site (OATS). Similar to the traditional three antenna method, three pairings of transmission measurements are taken. In each measurement, one antenna is set at a fixed height above the ground plane, while the other antenna is scanned in height over 1 to 4 m heights. Magnitude and phase responses between the two antennas are taken at multiple heights. Measured results are fit to a theoretical model using a complex fit algorithm. From this process, the gain and frequency dependent phase center locations of each antenna can be solved. Measurement data show that it is effective in reducing systematic uncertainties associated with assuming fixed phase center locations. In addition, unlike other calibration methods over a conducting ground plane, no assumptions are made about the antenna patterns. This method provides an accurate, versatile and fast method for calibrating LPDAs from as low as 100 MHz.
Antenna, Radome, and RCS testing systems rely on high-fidelity positioner systems to provide high-precision positioning of articles for RF testing. Historically, the industry has relied on linear PID control techniques in torque, velocity, and position control loops on individual axes to drive the positioners. Recently, advancements have been made in the use of advanced control hardware including multiple-DOF laser and optical feedback devices, brushless DC motors, VFD AC motors, and multi-drive torque-biased actuation. Advanced control techniques including single-axis error correction, multi-axis global error compensation, and multi-axis coordinated motion have been implemented to improve positioner accuracy. Here, a survey is conducted of control technologies in other industries such as machine tools and industrial robotics. An assessment is conducted on the viability of other advanced techniques to provide insight into the potential future control and capabilities of positioning systems in the RF testing industry. Candidate advanced techniques include gain scheduling and sliding-mode control which could provide improved accuracy over a wider range of conditions including varying loads and operating points caused by differing movement speeds or large variations in static loading. Dynamic input-shaping and feed-forward techniques could help suppress dynamic vibrations and improve dynamic tracking behavior for improved continuous-measurement scanning accuracy. Adaptive and non-linear control techniques might improve disturbance and error rejection for improved accuracy while managing dynamic-behavior drift allowing for adaptation to long-term positioner changes without re-tuning.
RCS Measurements at the upper half of the VHF range of the spectrum have become increasingly important. This type of measurement is usually performed in an outdoor RCS range. The present paper shows a design for an antenna that can be used to illuminate a reflector or as the illuminating structure in a RCS measurement. The antenna is fairly compact given the wavelength and exhibits a low VSWR and a good time domain performance for use with pulses. The new antenna has low side-lobes that otherwise could illuminate adjacent structures to the outdoor range and reduce the dynamic range of the measurements, this is an improvement over the Open-Boundary Quad-Ridged Horns Introduced over the past 9 years. The new Feed is a Quasi-Open-Boundary Horn, in which RF absorber material is used to create the Open Boundary behavior, but an enclosed structure is used to block the potential side-lobe radiation.
Abstract— The National Institute of Standards and Technology (NIST) presents a plan for evaluating and verifying the performance of the refurbished Ft. Huachuca Antenna Test Facility outdoor compact range. This plan was drawn up based upon information supplied to NIST on the intended applications.
The technique is validated with a 14-element printed circuit 2-12 GHz log-periodic antenna with known phase center, and applied to a 4-element printed circuit 2.4 GHz Yagi-Uda antenna with unknown phase center. A conventional phase center determination based on phase curvature and the proposed technique yield substantially the same results. It is shown that the phase center of a Yagi antenna is not located at the feed but is located closer to the center of the physical array.
Abstract—It has been shown that it is possible to get a good estimation of the location of the largest centers of reflection causing ripple in the quiet zone using spherical near-field scanning of the quiet zone in combination with back projection to far-field. This method however, suffers from poor resolution at lower frequencies making it hard to distinguish small contributions from the main beam if they are closely spaced. For this purpose the CLEAN algorithm has been adapted and is presented here.
I. INTRUDUCTION In the heating process of microwave absorbers under incident electromagnetic waves, two disciplines of physics are intertwined, i.e., electromagnetic waves behavior governed by Maxwell’s equations and heat transfer process dictated by laws of thermodynamics. The power density in the absorbers due to the electromagnetic .eld is given by p= s|E|2 =2po0 o ' f|E|2 (1) where, E is the total electric .eld (V/m) in the material, s is electrical conductivity of the material (S/m), o0 is the free space permittivity (8.854 × 10-12 F/m), o' is the imaginary part of the relative dielectric constant, and f is the frequency in Hz. This is point form of the Joule’s law, and is well understood by RF engineers. The EM behavior of the polyurethane absorbers can be numerically computed. The EM .eld acts as the heating source, and its distribution in the absorber can provide a good indication on the locations of hot spots. Polyurethane foam is an excellent insulator, so the conductive heat loss may be minimal. The heat exchanges can be reasonably described by radiation and convection transfers. Radiation takes place in the form of EM wave, mainly in the infrared region. The net power transferred from a body to the surroundings is described by Stefan-Boltzmann’s law [1], prad = osA(T4 -T04 ) (2) where A is the surface area, T is the surface temperature of the radiation body in K, and T0 is the ambient temperature in K. Unfortunately, the conventional symbols used in heat transfer s and o are not the same as those in Eq. (1). s here is the emissivity or emission coef.cient, and is de.ned as the ratio of the actual radiation emitted and the radiation that would be from a black body. o in Eq. (2) is the Stefan-Boltzmann constant (5.67 × 10-8 W/m2 K4 ). The context in the paper should make it clear which symbols the authors are referring to. Otherwise, we will make explicit references. The convective heat transfer is due to the motion of air surrounding the absorbers. Two forms can take place, naturally or by forced air. The relationship is described by Newton’s law of cooling [1]: pconv = hA(T -T0 ) (3) where h is the convection heat transfer coef.cient in (W/m-2 K-1 ). h is often treated as a constant, although it can be a function of the temperature. Eq. (3) assumes that the ambient air is abundant, and is taken to be constant. This is a reasonable assumption, because the heating is typically con.ned to a small localized area in a relatively large anechoic chamber. Combining the two mechanisms of heat transfer, the total heat loss is given by p= osA(T4 -T4 )+ hA(T -T0 ) (4) 0 It is possible to solve for the temperatures from coupled Maxwell’s and heat transfer equations. Realistic results require accurate electrical and thermal properties of the materials. It is often a non-trivial process to obtain the material properties in and of itself. Careful validation is warranted before we can have full con.dence in the results. In this paper, we adopt a measurement approach instead. We conduct a series of experiments to measure the temperature both on the surface of the absorbers using an infrared imaging camera, and internally using thermocouple probes inserted into the absorbers. Temperature pro.les versus applied E .eld are experimentally established. From the measured data, we curve .t to Eq. (4) or other mathematical functions. These functions are useful to calculate results at other .eld levels, e.g., extrapolating to a higher .eld where measurement results cannot be readily obtained. II. FIELD DISTRIBUTION INSIDE THE ABSORBERS Numerical analysis was performed using Ansys HFSS, a commercially available Finite Elements software package. As it was described in [2], symmetry is taken advantage of, so only one quarter of the pyramidal absorber is solved. The quarter pyramid is located inside a square cross section prism that bounds the computational domain. The structure is fed using a port located on the top of the geometry and the side boundaries of the domain are set as perfect electric conductor (PEC) or perfect magnetic conductor (PMC). The base is modeled as PEC. This is exactly the same approach taken in [2]. The structure of a CRV-23PCL-4 is analyzed at 12.4 GHz, the same frequency as used in the measurements. The resulting .eld is extracted at one plane. The plane is one of the two orthogonal planes that cut the pyramid in 4 sections. Fig. 1 shows the .eld distribution at 12.4 GHz. The curvature of the absorber pro.le has been added for clarity. The results are an approximation. The permittivity of the material is assumed to be fairly constant from 6 GHz to 12 GHz. The purpose of the numerical analysis is to check the expected .eld distribution in the pyramid, which we can use to compare with the infrared (IR) images of the absorbers taken during the measurements. Fig. 1. Electric Field distribution at 12.4 GHz The .eld distribution data shows that most of the .eld exists on the upper third of the pyramid. It also shows that there is a region of high .eld existing in the valleys between the pyramids. The surface temperature pro.le from the IR pictures shows that this is an real phenomena. On the other hand, the .eld is higher at the very tip of the absorber. Measurements from the IR images seem to contradict this result. This can be explained. Since the tip is smaller, it cools faster to the surrounding ambient temperature. III. EXPERIMENTAL SETUP AND DATA Experiments were performed on ETS-Lindgren CRV-23PCL-8, and CRV-23PCL-4 absorbers at 12.4 GHz. Both types are 23” long from tips to bases. A piece has a base size of 2’ × 2’. A CRV-23PCL-8 piece consists of 8×8=64 pyramids, whereas a CRV-23PCL-4 piece consists of 4×4=16 pyramids. The two types are designed to have similar RF performances, but the CRV-23PCL-8 is made of slender pyramids to facilitate better heat transfers to the surroundings [2]. The absorbers are mounted on a particle board with metallic backings, and are placed in front a Ku band horn antenna with a circular aperture (the gain is approximately 20 dBi). A 300W ampli.er is used, and the power to the antenna is monitored through a 40 dB directional coupler connected to a power meter. The test setup is shown in Fig. 2. The ambient temperature is at 23.C. Fig. 2. Test setup using a conical horn antenna to illuminate the absorbers As a .rst step, a 200 V/m .eld is generated by leveling to a calibrated electric .eld probe. The distance from the probe to the antenna is 30”. At this distance, near .eld coupling is assumed neglegible, and the incident wave uniform (numerical simulation also validated these assumptions). The power needed to generate 200 V/m .eld is recorded. Next, the .eld probe is replaced with the absorbers under test. The tips of the absorbers are placed at the same distance (30”) from the antenna. Other .eld strengths can be leveled by scaling from the power for 200 V/m. A. Surface Temperature Figs. 3 and 4 show two examples of the infrared images taken after the temperature reached equilibrium under a constant 700 V/m CW at f=12.4 GHz for the two types of absorbers described earlier. There is no forced air.ow during the measurement. Table 1 summarizes the resulting temperatures on the absorber surfaces at different .eld levels. Tests were performed on two .nishes of otherwise identical CRV-23PCL-8 absorbers, i.e., fully covered with rubberized paint, or with latex paint. The data indicates that the paint has minimal effects on absorber temperatures. Table 1 also lists data for the wider CRV-23PCL-4 absorbers (with latex paint). B. Internal Temperature of the Absorber recorded by Thermocouples Three thermocouples are inserted in the CRV-23PCL-8 which are painted with rubberized coating. They are inserted at distances of 4”, 6”, and 8” from the tip of the pyramid, as illustrated in Fig. 5. Fig. 6 shows the temperatures measured by the three sensors. The temperatures at 8” from the tip are consistently higher than at other locations. There is a gap in the data at 700 V/m because RF power was turned off brie.y. Internal temperature reached 115 .C under 1.7 kW/m2 Fig. 3. Infrared camera image for incident electric .eld of 700 V/m. The absorber is the slender CRV-23PCL-8. Fig. 4. Infrared camera image for incident electric .eld of 700 V/m. The absorber is the wider CRV-23PCL-4. (800 V/m). Since the maximum allowed temperature for the polyurethane foam material is 125 .C, the incident power density is recommended to stay less than 1.7 kW/m2 for CRV-23PCL-8 absorbers mounted vertically and with natural convection in a 23.C room. After the temperature reached equilibrium under 800 V/m, additional air.ow was introduced by turning on a 6” diameter fan at 45” in front of the absorbers. The air.ow rate was measured to be approximately 80 ft/min at this distance. Note that this is a rather moderate air.ow, which can arise naturally from air-conditioning vents in a chamber. As shown in Fig. 6, the internal temperature quickly dropped to 102.C from 115.C. TABLE I MAXIMUM SURFACE TEMPERATURE RECORDED BY THE IR CAMERA (AT EQUILIBRIUM). T0 =23. C. E Power CRV-23PCL-8 CRV-23PCL-8 CRV-23PCL-4 (V/m) Density rubberized latex (. C) rubberized (kW/m2 ) (. C) (. C) 200 0.11 24 300 0.24 28 360 0.34 30 400 0.42 35 36 43 500 0.66 41 50 600 0.95 54 67 700 1.30 63 82
In this paper we describe the NGS calibration facility and calibration hardware, and discuss the motivation for providing calibration services. We provide the time-difference, single-difference carrier phase observable models and estimation strategy currently used to generate NGS absolute calibrations. Calibration examples are provided in the accompanying poster presentation.
Abstract—In this paper, a novel feeding method of microstrip line array is presented, with which every line array can be fed from the side part instead of from the center part .At the same time a novel compact slot patch antenna loaded with a pair of spirals is proposed, which is 66% smaller than the conventional half-wave patch antenna. The simulated and measured radiation patterns at center frequency are both presented. The -18dB side-lobe level for the sum pattern and -37dB null depth for the difference pattern have achieved in the experiment.
I.
In the literature, one can find a low scattering photodiode modulated-probe for microwave near field imaging. The frequency response of the probe is computed at 2.45 GHz. In this paper, however a new formulation for computing the scattered field for low frequencies (from 150 MHz) by a broadband near field probe based on the impedance matrix is developed. In addition, a method to increase the scattered power by controlling matching will be shown.
Key words: 900 Microstrip Coupler, Metamaterial, Size Reduction, Split Ring Resonators, Rose Curve Resonators
Abstract—Cognitive Communications techniques are important for white space communications and spectrum sharing. Disaster recovery and green field communications could also be aided by such Context Aware systems. These networks could benefit from new compact, multi-band, active antenna implementations.
MI Technologies has delivered two new state of the art compact range measurement systems and has worked with Panasonic to develop automated test systems that have reduced the test time by more than a factor of 4. The range design includes significant automation, integration with the antenna’s built in up and down converters, and the ranges are reversible.
Implementation of a Novel Low-Cost Low-Profile Ku-Band Antenna Array for Single Beam Steering from Space TMM 4TMM 13i25mil 50mil 150mil 5° g Nicholas K. Host, Chi-Chih Chen, John L. Volakis