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Nowadays many antennas are planar and processed on single or multilayer PCBs. These antennas are often simulated before with a full-wave solver like CST Microwave Studio. In these simulations the reflection coefficient on the feeding transmission line is determined (in parallel to the farfield characteristics but which is of no interest in this paper). The antenna is optimized for minimal reflection. In the simulation the parameters of the antenna feeding line is chosen in the same manner than in the practical design where the antenna is directly connected to the output of an active circuitry via a planar transmission line (e.g. microstrip line, coplanar line, etc.). It is recommended to measure the input reflection coefficient of the antenna with a network analyzer before using it in the final design. For this reason the antenna is processed as prototype on a PCB and linked to the measurement equipment via an RF connector, eg. a Rosenberger 32K243 connector. It is common knowledge that every connector has an influence on signal transmission and provokes reflections. Thus the combination of antenna with connector can cause non-negligible reflections although the sole antenna has not. In order to determine the correct reflection coefficient of the antenna the transfer function of the connector must be known. In two former papers the authors presented methods to determine the scattering parameters of connectors by narrowband and broadband measurement of symmetrical structures. By knowing the scattering parameters the antenne input reflection coefficient can be determined by deembedding. Nevertheless the antenna is optimized for the unembedded design and not for the embedded version with connector. If the connector has non-neligible reflections in the high frequency domain the performance of the complete antenna prototype is bad and could not be used for other purposes (e.g. modular electronic setup where different components are linked together via cables). Furthermore if less power is transmitted to the PCB the calculation accuracy for the deembedding is reduced. A matching network can help to reduce the input reflection loss towards the antenna prototyp which is in general inserted in front of a device. A ”self-made” passive matching network between connector and coaxial cable at 24 GHz again requires a structure on a PCB which again introduces two cable to PCB transitions and additional losses. Thus the idea is to develop a matching on the PCB-side ”behind” the connector which reduces the input reflection of the connector independently of the termination structure on the PCB.
Radar signature measurements of targets with or without camouflage in different backgrounds using airborne SAR is complicated and expensive. Measurements at many orientations as well as illumination angles have to be performed for each target for completeness. A more efficient solution is to use ground based ISAR measurements of the desired targets and then blend these images into measured SAR scenes. We are developing a SAR-ISAR blending method where the target and background are modelled by point scatterer representations. This can be formulated as an inverse problem described by the equation Ax = y, (1) where A is a forward operator describing the model, x is the image and y is the measured RCS data. The point scatterer representations for the target and the SAR background are determined by solving (1). The main contribution of this paper is that we use a combination of L1 and L2 regularization methods to solve the inverse problem. The target measured by ISAR is sparse in the image domain and (1) is therefore solved efficiently using a L1 regularization method. However, the SAR background is not sparse in the image domain and (1) is therefore solved using a L2 regularization method. We use the following procedure: Define the operators A and At, where At is the conjugate transpose of A. The same operators are used for both the target and the background. Solve (1) using L1 regularization for the target measured using ISAR. Edit the target point scatterers so that only target related scatterers are included. Solve (1) using L2 regularization for the SAR background. Edit the background point scatterers by removing the shadowed region, alternatively attenuate if there is a camouflage net. Combine the edited point scatterers for the target and background and calculate the RCS for the combination. Add estimated system noise. Create a blended SAR-image. The method is demonstrated with ISAR measurements of a full-scale target, with and without camouflage, signature extraction and blending into a SAR background. We find that the method provides an efficient way of evaluating measured target signatures in measured backgrounds.
At NIST, we have developed a precision, wide-band, mmWave modulated-signal source with traceability to primary standards. We are now extending the traceability path for this modulated-signal source into free space to be used for verifying over-the-air measurements in 5G, wireless receivers. However, to obtain a traceable modulated signal in free space, the full scattering matrix of the radiating antenna must be measured. We have extended the extrapolation methods used at NIST, based on the work of Newell, et al. [1]. The extrapolation measurement provides a very accurate, far-field, on-axis, scattering matrix between two antennas. When combined with scattering-matrix measurements made with permutations of pairs of three antennas, far-field scattering, and, thus, gain, is obtained for each antenna. This allows an accurate extrapolation of the antenna’s near-field pattern. We have incorporated the extrapolation fitting algorithms into a Monte Carlo uncertainty engine called the NIST Microwave Uncertainty Framework (MUF) [2]. The MUF provides a framework to cascade scattering matrices from various elements, while propagating uncertainties and maintaining any associated correlations. By incorporating the extrapolation measurements, and the three-antenna method into the MUF, we may provide traceability of all measurement associated with the gain, including the scattering parameters. In this process, we studied several aspects of the gain determination. In this work, we show simulations determining the efficacy of filtering to reduce the effect of multiple reflection on the extrapolation fits. We also show comparisons of using only amplitude (as is traditionally done) to using the full complex data to determine gain. Finally, we compare uncertainties associated with choices in the number of expansion terms, systematic alignment errors, uncertainties in vector network analyzer calibrations and measurements, and phase error introduced by cable movement. With these error mechanisms and their respective correlations, we illustrate the NIST MUF analysis of the antenna scattering-matrix with data at 118 GHz. [1] A. C. Newell, R. C. Baird, and P. Wacker “Accurate Measurement of Antenna Gain and Polarization at reduced distances by an extrapolation technique” IEEE Transactions on Antennas and Propagation. Vol. 21, No 4, July 1973 pp. 418-431. [2] D. F. Williams, NIST Microwave Uncertainty Framework, Beta Version. NIST, Boulder, CO, USA, Jun. 2014. [Online]. Available: http://www.nist.gov/pml/electromagnetics/related-software.cfm
Millimeter-wave scanners are a powerful tool in multiple fields such as security and non-destructive evaluation. Recent advances in design and manufacturing at this frequency band, also boosted by automotive and communication industries, are resulting in the first generation of portable scanners based on different imaging paradigms. In the recent work of the authors, the capabilities of this kind of portable scanners have been considered. In particular the evaluation of different methods to combine multiple (potentially overlapped) acquisitions from arbitrary points has been considered [1]. The proposed imaging method for each local acquisition was based on Synthetic Aperture Radar (SAR) techniques. Nevertheless, this kind of imaging method usually requires a dense sampling and, consequently, it can result in a large number of transceivers increasing the cost and weight of the device. For this reason, the use of multistatic arquitectures, also known as multiple-input multiple-output (MIMO), similar to the ones proposed in [2] is considered in this communication. This approach enables to reduce the total number of elements by almost an order of magnitude by placing transmitters and receivers at different positions along the scanner aperture in contrast to conventional SAR that considers a dense aperture of equally spaced independent transceivers. As demonstrated in [2], if the position of transmitters and receivers is properly designed, the obtained results are equivalent to the ones provided by a conventional dense sampling. [1] Jaime Laviada, Yuri Álvarez, Ana Arboleya, Fernando Las-Heras, Borja González Valdés, “Multiview Techniques for mm-Wave Imaging” Antennas and Propagation Simposium AP-S 2017, San Diego, USA, 2017. [2] S. Ahmed, A. Schiessl, and L. Schmidt, “A novel fully electronic active real-time imager based on a planar multistatic sparse array,” IEEE Transactions on Microwave Theory and Techniques, vol. 59, no. 12, pp. 3567–3576, 2011.
Abstract— Indoor antenna ranges must have the walls, floor and ceiling treated with RF absorber. The normal incidence performance of the absorber is usually provided by the manufacturers of the materials; however, the bi-static or off angle performance must also be known. In reference [1], a polynomial approximation was introduced that gave a prediction of the reflected energy from pyramidal absorber. In this paper, the approximations are used to predict the quiet zone (QZ) performance of several anechoic chambers. These predictions are compared with full wave analysis performed in CST Suite®. A 12 m wide by 22 m long with a height of 12 m chamber was analyzed at 700 MHz. The QZ performance was compared to the polynomial predictions showing a difference of less than 2.2 dB. In addition, comparisons are made with measurements of the QZ performance of anechoic chambers. Measurements performed per the free space VSWR method of three different chambers are compared with the prediction that uses the polynomials presented in [1]. The chambers are: a 18 m long by 11.5 m wide and 11.5 m in height operating from 100M MHz to 12 GHz; a 13.41 m by6.1 m by 6.1 m operating from 800 MHz to 6 GHz; and a 14 m long by 4.12 m by 4.27 m operating in the X band. The results show that the polynomial approximations can be used to give a reasonably accurate and safe prediction of the QZ performance of anechoic chambers. [1] V. Rodriguez and E. Barry, “A polynomial approximation for the Prediciotn of Reflected Energy from Pyramidal RF Absorbers,” Proceedeings of the 38th annual Symposium of the Antenna Measurement Techniques Association (AMTA 2016), pp. 155–160, October 2016.
Abstract— RF shielded enclosures have been common features in laboratories and manufacturing areas for over 70 years. They provide a quiet environment where RF measurements can be performed without interference from outdoor sources and are used to keep potentially classified frequencies and modulations from leaking out. In general, these shielded rooms have shielded doors to maintain the shielding integrity of the enclosure until they are opened. In some cases, to maintain the shielding integrity as personnel moves from the inside to the outside of the room and vice-versa, dual shielded doors with a small vestibule between them are used. However, the presence of multiple doors increases the time to access the enclosure. To solve this, some enclosures are designed featuring access passages to maintain the shielding integrity over a broad frequency without the use of doors. Although this type of access has been around for over 40 years, its design has never been discussed in the literature. In this paper, a door-less access is analyzed and some design rules are presented. The limitations of these accesses are also presented. While clearly they do not have the shielding performance of a shielded door, they are ideal for certain applications. Keywords: Shielded enclosures, door-less access
Abstract— Tilting the receive end wall of a compact range anechoic chamber to improve Radar Cross-Section (RCS) measurements has been a tool of the trade used since the earliest days of anechoic chambers. A preliminary analysis using geometrical optics (GO) validates this technique. The GO approach however ignores the backscattering modes from the reflected waves from a field of absorber. In this paper, a series of numerical experiments are performed comparing a straight wall and a tilted wall to show the effects on both the quiet zone and the energy reflected back towards the source antenna. Two Absorber covered walls are simulated. Both walls are illuminated with a standard gain horn (SGH). The effects of a wall tilted back 20° are computed. The simulations are done for 72-inch long absorber for the frequency range covering from 500 MHz to 1 GHz. The ripple on a 10 ft (3.05 m) quiet zone (QZ) is measured for the vertical wall and the tilted wall. In addition to the QZ analysis a time-domain analysis is performed. The reflected pulse at the excitation antenna is compared for the two back wall configurations Results show that tilting the wall improves measurements at some frequencies but causes a higher return at other frequencies; indicating this method does not provide a broadband advantage. Keywords: Anechoic Chamber Design, Radar Cross Section Measurements, Geometrical Optics
Equivalent Isotropically Radiated Power (EIRP), Saturating Flux Density (SFD) and Group Delay (GD) are three system level parameters often measured during the characterization of spacecraft systems. EIRP is of interest for transmitters, SFD for receivers and GD for the entire up/down link. A test methodology for EIRP and SFD was first presented in [1] and [2] and a detailed procedure presented in [3]. To date GD has only been measured under far-field (or simulated far-field) conditions. In [4], a concept for measuring GD in a planar near-field (PNF) range is described, but no methodology is presented. In this paper, we present a method for measuring GD in a planar near-field range. The technique is based on a set of three antenna pairs, measured sequentially, from which the insertion phase of the measurement system and the near-field probe [5] can be resolved. Once these parameters are known, insertion phase for the device under test (i.e. a Tx or Rx antenna) can be measured and GD calculated as the negative frequency derivative of the insertion phase. An added complexity in the case of a near-field measurement is the near-field probe is in close proximity to the device under test (not far-field condition) for which compensation is needed. We show through simulation and measurement, that the plane wave expansion allows us to compute a correction factor for the proximity of the probe to the device under test; thus allowing correction of the measured insertion phase. The final step in measuring payload GD through both uplink and downlink channels is to set up a fixed Tx probe in close proximity to the Rx antenna and an equivalent Rx probe in close proximity to the Tx antenna and performing a through measurement as one would do on a far-field range. Correction factors for compensating for the proximity of both probes are then applied, based on independent a-priori Rx and Tx case measurements performed on the antennas. Simulated and measured data will be presented to demonstrate the process and to illuminate some of the finer nuances of the correction being applied. Index Terms— Group Delay, Planar Near-Field, Antenna Measurements, Three Antenna Method. [1] A. C. Newell, R. D. Ward and E. J. McFarlane, “Gain and power parameter measurements using planar near-field techniques”, IEEE Trans. Antennas &Propagat, Vol 36, No. 6, June 1988 [2] A. C. Newell, “Planar near-field antenna measurements”, NIST EM Fields Division Report, Boulder, CO, March 1994. [3] D. Janse van Rensburg and K. Haner, “EIRP & SFD Measurement methodology for planar near-field antenna ranges”, Antenna Measurement Techniques Association Conference, October 2014. [4] C. H. Schmidt, J. Migl, A. Geise and H. Steiner, “Comparison of payload applications in near field and compact range facilities”, Antenna Measurement Techniques Association Conference, October 2015. [5] A. Frandsen, D. W. Hess, S. Pivnenko and O. Breinbjerg, “An augmented three-antenna probe calibration technique for measuring probe insertion phase”, Antenna Measurement Techniques Association Conference, October 2003.
Microstrip patch antennas are well known in the field of communications and other areas where antennas are used. They consist of a metallic conducting surface deposited onto a grounded dielectric substrate and are widely used in situations where a conformal antenna is desired. They are also popular antennas for array applications. But most patch antennas are typically resonant structures owing to the standing wave of current that forms on them. This resonant behavior limits the impedance bandwidth of the antenna to a few percent. In this paper we shall present an approach for improving the bandwidth of a resonant patch antenna which employs an engineered anisotropic superstrate. By proper design of this superstrate and its tensor, and proper alignment of it with the axis of the patch, an antenna with improved impedance bandwidth results. Some of the challenges associated with the measurement of the anisotropic superstrate will be discussed, ranging from 3D simulations to physical models tested in the laboratory. A final working model of the antenna will be discussed; this model consists of a stacked patch arrangement and was designed to operate at the GPS L1 and L2 frequencies. Data collected from 3D simulations using CST Microwave Studio along with laboratory and anechoic chamber measurements will be presented, showing how the bandwidth at both of these frequencies can be increased while maintaining circular polarization in both passbands. Tolerance to errors in alignment and fabrication will also be presented. Additionally, some lessons learned on anechoic chamber measurements of the antenna’s gain and axial ratio will be discussed.
In June of this year, DSC completed the installation of a turnkey RCS measurement system that is used for in-process verification (IPV) and final component validation using standard near field QC techniques in an echoic chamber. The delivered system included a radar, antennas, shroud, ogive pylon, foam column, elevators for each – column and pylon, automated pit covers, test bodies, target transport carts, and calibration targets. The system automatically loads test objects on the correct target support system, requiring no action by the operator to connect a target onto the azimuth over elevation “tophat” positioner – it is all automatic. The user interface is designed to be operated by production line workers, greatly reducing the need for experienced RCS test engineers. Simple pass/fail indicators are shown to the test technicians, while a full detailed data set is stored for engineering review and analysis. A wall display guides users through a test sequence for target handling and starting the radar. Radar data collection of all azimuth and elevation angles and target motion are initiated from a single button push. This is followed by all data processing necessary to conduct the ATP on the parts providing a pass/fail report on dozens of parameters. The application of production line quality automation to RCS measurements improves the repeatability of the measurements, greatly reduces both measurement time as well as overhead time, and allows systems operators to become more interchangeable. This highly successful project, which was completed on-time and on-budget, will be discussed. This discussion will include radar performance, antenna and shroud design, target handling, data processing and analysis software, and the control system that automates all the functions that are required for RCS measurements.
Measurement of the radiation properties of low gain antenna operating at VHF frequencies is well known to be a challenging task. Such antennas are sometimes tested in outdoor Far Field (FF) ranges which are unfortunately subject to errors caused by the electromagnetic pollution and scattering from the environment. Near Field (NF) measurements performed in shielded anechoic chambers are thus preferable to outdoor ranges. However, also in such cases, the accuracy of the results may be compromised by the poor reflectivity of the absorbing material which might be not large enough wrt the VHF wavelength. Other source of errors may be caused by the truncation of the scanning area which generates ripple on the FF pattern after NF/FF transformation. Spherical multi-probe systems developed by MVG are optimal measurement solution for low directive Device Under Test (DUT). Such systems allow to perform a quasi-full spherical acquisition combining a rotation of the DUT along azimuth, with a fast electronically scanned multi-probe vertical arch. The DUT can be accommodated on masts made of polyester material which allows to minimize the interaction with the DUT. Measurements of low directive device above 400 MHz performed with such type of systems have been demonstrated to be accurate and extremely fast in previous publications. In this paper, measurements of a low directivity antenna, performed at VHF frequencies in a MVG spherical multi-probe system, will be presented. The antenna in this study is an array element, part of a larger array, which has been developed for space-born AIS applications. Gain and pattern accuracy of the measurement will be demonstrated by comparison with full wave simulation of the tested antenna.
This paper demonstrates two known analog methods of boosting the receive power gain of a wireless communication link inside a reflective environment for applications inside aircraft cabins and other cavities in support of wireless avionics intra-communication (WAIC) technologies and standards. The first method consists of using multiple antennas (antenna diversity), and the second consists of using multiple frequencies (frequency diversity). A series of tests were conducted inside an RF reverberation chamber. RF absorbing material was introduced into the chamber to match the loading characteristics of a Boeing 727 aircraft as reported in literature. To demonstrate the benefits of antenna diversity, a 3 GHz continuous wave signal was injected into the chamber using an RF source and three omni-directional antennas were used to capture the same source inside the chamber. The amplitude of the power fluctuation due to multi-path at the receive antennas was recorded using three separate spectrum analyzers. The chamber’s mechanical stirrer rotated continuously while the amplitude time recordings were made. Frequency diversity advantages are similarly demonstrated by transmitting from a vector signal generator two continuous wave frequencies at 2.99 GHz and 3.01 GHz out of a single transmit antenna. Two separate spectrum analyzers record the two frequencies from a single receive antenna. Cumulative distribution functions are computed from which diversity gains are determined. Both methods resulted in substantial received power gain over the single transmit and single receive method. In addition, measurement of wireless parameters such as Rician K-factor and delay spread at various antenna separation distances are illustrated using a basic network analyzer setup. The preliminary demonstrations show that a reverberation chamber could useful for testing of future WAIC protocols and effective as a tool to mitigate some of the risks inherent to wireless system for critical functions.
This paper presents the results of a new measurement technique to determine antenna system noise temperature using data acquired from a planar near-field measurement. The ratio of antenna gain to system noise temperature (G/T) is usually determined in a single measurement when the antenna is alternately pointed towards the “cold sky” and a hot radio source such as the sun or a star with a known flux density. The antenna gain is routinely determined from near-field measurements and with the development of this new technique, the system noise temperature can also be determined. The ratio of G/T can therefore be determined from planar near-field data without moving the antenna to an outdoor range. The noise temperature is obtained by using the plane-wave spectrum of the planar near-field data and focusing on the portion of the spectrum in the evanescent or “imaginary space” portion of the spectrum. Near-field data is obtained using a data point spacing of l/4 or smaller and the plane-wave spectrum is calculated without applying any probe correction or Cos(q) factor. The spectrum is calculated over real space corresponding to propagating modes of the far-field pattern and also the evanescent or imaginary space region where . Actual evanescent modes are highly attenuated in the latter region and therefore the spectrum in this region must be produced by “errors” in the measured data. Some error sources such as multiple reflections will produce distinct localized lobes in the evanescent region and these are recognized and correctly identified by using a data point spacing of less than l/2 to avoid aliasing errors in the far-field pattern. It has been observed that the plane wave spectrum beyond these localized lobes becomes random with a uniform average power. This region of the spectrum must be produced by random noise in the near-field data that is produced by all sources of thermal noise in the electronics and radiated noise sources received by the antenna. By analysing and calibrating this portion of the spectrum in the evanescent region the near-field noise power can be deduced and the corresponding noise temperature determined. Simulated and measured data will be presented to illustrate and validate the measurement and analysis techniques. Keywords — Planar Near-Field, G/T, Figure-of-Merit Measurements, Simulation, Plane Wave Spectrum.
The topic of non-redundant near-field sampling has received much attention in recent literature. However, a practical implementation has so far been elusive. This paper describes a first step toward such a practical implementation, where the practicality and generality are maximized at the expense of more acquired data points. Building on the theoretical work of faculty at the University of Salerno and University of Naples, the authors have acquired a set of near-field data using a spiral locus of sample points and, from those data, obtained the far-field patterns. In this paper, we discuss the acquisition system, the calculation and practical implementation of the spiral, the phase transformations, interpolations, and far-field transforms. We also present the resultant far-field patterns and compare them to patterns of the same antenna using conventional near-field scanning. Qualitative results involving aperture back-projection are also given. We summarize our findings with a discussion of error, uncertainty, acquisition time, and processing time in this simplified approach to non-redundant sampling in a practical system.
In millimeter wave near-field measurements dual polarized probe system can be used with some of the advantages: the two electric field components are simultaneously measured within a single scan, amplitude and phase drift affects the two polarization components in the same way and there is no need of mechanical rotation of the probe. Today at DTU-ESA Facility we have dual-polarized probes in range 400MHz-40GHz and this study is part of extending the operational frequency range of the DTU-ESA Facility up to 60GHz. First order µ = ± 1 rotationally symmetric probes are desired because they employ an efficient data-processing and measurement scheme. In this work we design and test at DTU-ESA Facility a dual polarized first order probe system at 60GHz - a conical horn, including the elements: a pin diode SPDT (single pole double throw) switch up to 67GHz from Ducommun an OMT (ortho-mode transducer) from Sage Millimeter in 50-75GHz band with square waveguide antenna port (3.75mm) a square to circular transition (3.75mm to 3.58mm) from Sage Millimeter which is integrated between the OMT and conical horn 1.85mm connector cables up to 75GHz and two coaxial to waveguide adapters to connect the switch to the OMT from Flann Microwave To ensure accurate measurements at 60GHz, the hardware components were selected to provide a low cross polarization of the probe, the switch and the OMT having 40dB isolation between ports. The path loss at 60GHz is 83dB for a 6m distance and to compensate for such a loss, a 26dB gain is desired for the conical horn, which is simulated using WIPL-D software and in-house manufactured. The 60GHz dual-polarized probe is currently being assembled and will be tested in both planar and spherical near-field setups. In the full version of the paper calibration results will be shown but also results from using the probe as a probe for the measurement of a 60GHz AUT.
As modern antenna array systems for MIMO and 5G applications are deployed, there is increased demand for measurement techniques for timely calibration, at both research and commercial sites.[1] The desired measurement method must allow for the de-embedding of information about the closed digital signal chain and element alignment, and must be performed in the near-field. Current means of measuring large arrays cover a variety of methods. Single-element gain and pattern calibration must cover the parameter space of element weightings and is extremely time-consuming, to the point where the measurement may take longer than the duration over which the array response is stable.[4] Two other popular methods are the transmission of orthogonal codes and the use of holography to reconstruct a full-array pattern. The first of these methods again requires extremely long measurement time. For an array of N elements and weightings per element W_n, the matrix of orthogonal codes must be of an order greater than NW_n.[4][3]. This number varies with the form of W_n depending on whether the array is analog or digital, but in both cases for every desired beam configuration, an order-N encoding matrix must be used. The second method relies on illuminating subsets of elements within an array and reconstructing the full pattern.[2] Each illuminated subset, however, neglects some amount of coupling information inherent to the complete system, making this an imperfect method. In this work we explore the development of a sparse set of measurements for array calibration, relying on coherent multi-channel data acquisition of wideband signals at 75 GHz, and the hardware characterization and post-processing necessary to perform channel de-embedding at an elemental level for a 4x1 system. By characterizing the complete RF chain of our array and the differential skew and phase response of our measurement hardware, we identify crucial quantities for measuring closed commercial systems. Additionally, by combining these responses with precise elemental location information, we consider means of de-embedding elemental response and coupling effects that may be compared to conventional single-element calibration information and full-pattern array measurements. [1] C. Fulton, M. Yeary, D. Thompson, L. Lake, and A. Mitchell. Digital phased arrays Challenges and opportunities. Proceedings of the IEEE, 104(3):487–503, 2016. [2] E. N. Grossman, A. Luukanen, and A. J. Miller. Holographic microantenna array metrology. Proceedings of SPIE, Passive Millimeter-Wave Imaging Technology VIII, 5789(44), 2005. [3] E. Lier and M. Zemlyansky. Phased array calibration and characterization based on orthogonal coding Theory and experimental validation. 2010 IEEE International Symposium on Phased Array Systems and Technology (ARRAY), pages 271–278, 2010. [4] S. D. Silverstein. Application of orthogonal codes to the calibration of active phased array antennas for communication satellites. IEEE Transactions on Signal Processing, 45(1):206–218, 1997.
A complete characterization of the radiation and scattering phenomena is essential to the ray tracing simulators. In the ray tracing modeling, the electromagnetic field quantities are traced along the ray paths and determined by the antenna radiation pattern and the scattering patterns of the obstacles. The polarimetric patterns may be prepared in advance from the measurements or numerical simulations, and reused by the ray tracing simulators for various situations. However, the prefabricated pattern data set usually contain only a limited quantity of samples at discrete angular directions and frequencies. The lack of full representations of the desired patterns hinders the accurate calculation of ray field quantities. Although interpolation can be done using multidimensional splines or polynomials, the accuracy is not assured by the problems’ physics. Especially, it is difficult to tackle the phase wrapping problem in the multidimensional case, which might lead to wrong phase interpolation. In this paper, model based parameter estimation (MBPE) is used to circumvent the requirement of obtaining all samples of the desired radiation and scattering patterns in both spatial (angular) and frequency domains. Since any function defined on the surface of a sphere can be represented by a sum of spherical harmonic functions, we utilize the spherical harmonic expansion in the spatial domain firstly. Specifically, in order to avoid the singularities of conventional vector spherical harmonics at the north and south poles, scalar spherical harmonics is used instead. The expansion coefficients are vector-valued frequency domain responses, independent to the angular variables. Then each coefficient is expanded by using the singularity expansion method, which leads to a rational function characterized by its poles and residues in the frequency domain. Since the poles are the characteristic of the considered object (antenna or scatter), it is reasonable to assume that all spherical harmonic components have the same poles. Therefore, the parameters to be estimated are the frequency domain poles and the corresponding residues for each spherical harmonic component. By following this method, a physically-based, closed-form, reduced-order parametric model can be established from the sampled pattern data. The proposed method will be validated by simulations and measurements.
The maintenance of aircraft radomes is of particular importance for the commercial aviation industry due to the necessity to ensure the correct functioning of the radar antenna, housed within such protective enclosures. Given that the radar component provides weather assessment, as well as guidance and navigation functions (turbulence avoidance, efficiency of route planning in case of storms, etc.), it is imperative that every repaired radome be tested with accuracy and reliability to ensure that the enclosed weather radar continues to operate in accordance with the after-repair test requirements of the RTCA/DO-213. Recently, this quality standard was updated and published under the name RTCA/DO-213A, establishing more stringent measurement requirements and incorporating the possibility of measuring radomes using Near-Field systems. Consequently, a compliant multi-probe Near- Field system concept – AeroLab – has been specifically designed to measure commercial aircraft nose-radomes, in order to meet the new standard requirements. AeroLab performs Near-Field measurements. Near-Field to Far-Field transformations are then applied to the results. Such a Near-Field system allows the test range to be more compact than traditional Far-field test ranges, and thus be independent from the updated Far-Field distance which has progressed from D²/2l to 2D²/l in the new standard RTCA/DO-213A. AeroLab enables the evaluation of the transmission efficiency and beamwidth. It also allows for accurate evaluations of the side-lobe levels by providing improved visualization of principal cut views selected from 3D patterns. Moreover, depending upon the weather radar system inside the radome under test, 2 distinct scan sequences must now be taken into account: “elevation over azimuth” and “azimuth over elevation”. AeroLab emulates both of these motion sequences through a monolithic gimbal. Furthermore, thanks to its multi-probe array, such measurements are performed in a fraction of the time spent in current mono-probe test facilities (less than 4 hours, i.e. 1/3 less time than single probe scanners). Keywords: RTCA/DO-213A, radome measurement system, after-repair tests, multi-probe measurement system, Near-Field system.
For high speed and high data-rate communications, operating frequency bands of wireless communication systems have been moving to submillimeter frequency range and their bandwidths have been broadening. IEEE 802.15 THz Interest Group (IEEE 802.15 IGthz) has been performing a channel characteristics study for future indoor millimeter and submillimeter wireless communications in the frequency range of 75 - 110 GHz and 270 - 320 GHz. Specular reflectance data of indoor interior materials is a prerequisite to analysis of the channel characteristics of new indoor millimeter and submillimeter wireless communications. Specular reflectiondescribed by the law of reflection states that the direction of the incident wave and the direction of the reflected wave make the same angle with respect to the surface normal, thus theangle of incidence is equal to that of reflection. This paper describes a specular reflectance measurement system and shows measurement result of dielectric plates in the frequency range from 110 GHz to 325 GHz. Specular reflectance measurement system consists of an S-parameters measurement system and a specular reflectance measurement apparatus. The S-parameters measurement system consists of a 67 GHz vector network analyzer used as the main frame and three frequency extenders which are operating at three frequency bands (D-band (110 -170 GHz), G-band (140-220 GHz) and J-band (220-325 GHz)), respectively. The specular reflectance measurement apparatus consists of a transmitting part, a receiving part, and a MUT holder which is positioned in the middle of the transmitting and receiving parts. During the specular reflectance measurement, the transmitting part is fixed while the MUT holder and receiving part are coaxial-rotating with 1:2 speed ratio. The transmitting and receiving frequency extenders are installed on the transmitting and receiving parts, respectively. For the specular reflectance measurement, one measures the transmission coefficient (S21_MUT) corresponding to the specular reflectance of an MUT mounted on the MUT holder. After replacing the MUT with a metal plate, one measures the transmission coefficient (S21_metal) corresponding to the specular reflectance of the metal plate, assumed to be -1. Specular reflectance of the MUT is obtained by taking the ratio (S21_MUT/S21_metal) of the respective transmission coefficients corresponding to the specular reflectance of the MUT and the metal plate. Multiple reflection effects between the transmitting and receiving antennas can be averaged out and minimized by averaging the transmission coefficients measured with changing the separation distances between the two antennas by ?/8 interval (i.e. initial distance + n·?/8, n=0,1,2,3). Specular reflectances of dielectric plates are measured in the 30° to 70° incident angle range with the developed measurement system in the frequency range from 110 GHz to 325 GHz. Description of the detailed measurement system and measurement result will be presented at the symposium.
In this paper, we explore the possibility of using a photonics-based E-field sensor as a near-field probe. Relative to open-ended waveguide (OEWG) probes, a photonics probe could offer substantially larger bandwidths. In addition, since it outputs an optical signal, a photonics probe can offer signal transport through optical fiber with much lower loss than what can be achieved using RF cables. We begin with a discussion of the theory of the device followed by a summary of results of a photonics sensor that was tested in a spherical near-field (SNF) range. In these tests, data were collected with the photonics probe in the test antenna position to characterize various probe parameters including polarization discrimination, probe gain, effective dynamic range, and probe patterns. In the same set of tests, the photonics device was placed in the probe position in the range and used to measure patterns of two different antennas: a standard gain horn and a slotted waveguide array antenna. The resultant patterns are shown and compared to patterns collected with traditional RF probes. We conclude the paper with a discussion of some of the advantages and disadvantages of using a photonics probe in a practical system based on the lessons learned in the SNF testing.